1. Field of the Invention
The present invention relates generally to circuits for reducing and/or eliminating distortion in RF and microwave systems, and more particularly to a RF and microwave system linearizer using quadrature nonlinear distortion generators.
2. Discussion of Related Art Including Information Disclosed Under 37 CFR §§1.97, 1.98
The demand for higher information throughput in radio frequency (RF) and microwave communications systems such as advanced digital cellular phone systems has led to the use of modern, spectrally efficient modulation techniques such as WCDMA and FDM. Such spectrally efficient techniques however come with high waveform peak-to-average ratios (PAR). Circuits processing these signals must remain linear over this high instantaneous dynamic range to avoid generating nonlinear distortion that causes interference to other users. Maintaining linearity is especially a problem for the power amplifier (PA)—usually the most power-consumptive part of the system. Lowering PA average power (“power backoff”) to accommodate the peak powers and avoid nonlinear distortion also unacceptably diminishes its power efficiency. Efficiency and linearity are mutually exclusive in traditional PAs, and for good linearity with reasonable efficiency some form of linearization is needed [Vuolevi, J., Distortion in RF Power Amplifiers, Norwood, Mass.: Artech House, 2003]. Diminished efficiency worsens battery life in handhelds and worsens energy costs and thermal management issues in base stations. The common design paradigm has become to first design an efficient PA using one of the numerous available techniques [Raab, F. H, et. al., “Power Amplifiers and Transmitters for RF and Microwave,” IEEE Trans. Microwave Theory and Techniques, Vol. 50, No. 3, March 2002, pp. 814-826] and then invoke linearization to fix its linearity.
It needs to be noted that simply filtering out undesired nonlinear distortion products is ineffective as distortion products due to odd-degree nonlinearities manifest both within and very close to the fundamental frequencies. The nonlinearities within the fundamental frequency interfere with the in-band signals and worsens error vector magnitude (EVM), while nonlinearities close to the fundamental frequency cause adjacent channel power (ACP) that interferes with other users. Overall system capacity is diminished.
An important and difficult aspect of RF and microwave power amplifier linearization is the vector nature of the signals. At low frequencies distortion products are collinear with the fundamental signal, that is, they have a simple plus-or-minus sign relationship to the fundamental signal, either in-phase or antiphase. At RF and microwave frequencies, however, distortion phase can deviate from fundamental signal phase by arbitrary angles for many reasons, including reactive and physically remote nonlinearities. Designing for, and maintaining, the proper magnitude and phase relationships (that is, vector relationships) between distortion products and the ameliorative solution is a difficult aspect of linearization. Vector tuning of a signal typically comes in either polar format (magnitude and phase) or rectangular format (“In-phase” and “Quadrature”, or “IQ”). The polar format independently tunes magnitude and phase, but the phase modulation aspect is in general difficult to implement, especially if delay lines are used. The dynamical coordination of the magnitude and phase modulators is also very difficult, especially at the high modulation rates of modern systems. Further, inexpensive scalar spectrum analysis provides no indication as to whether magnitude or phase is preferably adjusted while tuning for lowered distortion, and so convergence can be very poor unless expensive vector spectrum analysis is invoked.
The physically large and mixed-technology aspects of many linearizer implementations preclude integration, thus increasing cost and environmental sensitivities. Physically large distributed elements such as delay lines are not uncommon [see, for example, U.S. Pat. No. 6,788,139, to Villemazet], but are limited to microwave frequencies, and their lengths are very difficult to adjust. Phase shifters are easier to tune but have the desired delay only at a single frequency, limiting instantaneous linearization bandwidth. Many linearization implementations also consume high DC power, in direct opposition to the primary engendering motivation. In broad terms, linearization techniques include careful circuit design, power backoff, feedback, feedforward, predistortion (both analog and digital), and derivative superposition.
Careful circuit design includes using quality components (particularly high linearity active devices), carefully chosen bias points and component values, and sound implementation. But this comes at the expense of design time, cost, and high DC power consumption. These procedures are well known and widely applied, so the drive for yet further linearity improvement requires the development of improved linearization techniques, an end to which the present invention is directed.
The power backoff approach reduces fundamental power levels to a small fraction of the standing voltage and current bias points, taking advantage of the even faster (relative to the fundamental) reduction rate of the distortion products. The diminished power efficiency quickly becomes unacceptable, however. Further, some forms of distortion found in Class B and deep Class AB amplifiers remain even at lower power levels [see, Steven Cripps, Advanced Techniques in RF Power Amplifier Design, Artech House, 2002, p. 79].
Systems using negative feedback at RF and microwave frequencies reduce fractional distortion by a factor of the loop gain [Sansen, W., “Distortion in Elementary Transistor Circuits,” IEEE Trans. Circuits and Systems—II, Analog and Digital Signal Processing, Vol. 46, No. 3, March 1999] and provide desensitivity to variations in forward path component values. But limited loop gains and excess phase shifts at such high frequencies forces severe and unacceptable tradeoffs amongst distortion suppression, bandwidth, and stability.
Envelope-rate negative feedback systems, such as Cartesian feedback, downconvert to baseband the modulated output signal for comparison to the baseband modulation inputs, taking advantage of the high loop gains available at the baseband frequencies. But errors in the feedback path, and in particular the added noise, delay, and nonlinearities of the downconverter, are not corrected by the loop and add directly to the signal. Additionally, sensitivity loop delays remains high on the scale of an RF period so as to make drift and instability a problem. The downconverter also adds appreciable cost and complexity to the system.
Feedforward systems generate an error signal (a scaled version of the added distortion of the PA) by subtracting the PA input and an appropriately scaled version of its output. The error signal is then appropriately amplified by an “error amplifier” and subtracted from the original amplifier output to cancel distortion. Feedforward systems have good linearity and very wide bandwidths. They are popular, despite their numerous disadvantages. They have an undesirable reliance on accurate tuning of amplitude scaling and time delays and, lacking negative feedback, they are sensitive to drifts in the forward-path component values. Adaptive systems are commonly needed to compensate for drift, adding further complexity and expense. Further, the power drawn by the error amplifier and the inefficiencies of the RF/microwave power combiner at the output serve to limit overall available efficiency.
Predistortion systems place a compensating nonlinearity in series with a distorting amplifier input. Postdistorters (in series with the output) are also possible, but predistorters are far more common as they operate at the lower power levels at an amplifier input and thus consume little DC power for better overall efficiency. Both analog and digital predistorters are common.
As with feedforward, predistorters are open loop and thus susceptible to drift, either in the amplifier being linearized (the “distorting main amplifier”, or DMA) or the linearizer itself. They are therefore commonly implemented within an adaptive system, so easy tunability is a desired feature. Predistorters are generally of different physical construction and in less-than-intimate contact with the DMA, exacerbating drift problems. Another issue with predistorters is that when canceling lower order distortion products, the cascade of predistorter plus distorting main amplifier generates new undesired distortion products of orders higher than those of the nonlinearities of either individual block [Steven Cripps, Advanced Techniques in RF Power Amplifier Design, Artech House, 2002, pg. 156],
Analog RF predistorters have the advantage of being inherently broadband because even very wide instantaneous modulation bandwidths constitute a small fraction of the RF center frequency. Their DC power consumption is therefore effectively independent of modulation bandwidth and they can even be entirely passive [Ibid., Ch. 5]. Digital predistorters (DPD), on the other hand, operate at baseband frequencies and must provide processing power (and thus consume DC power) directly proportional to the modulation bandwidth—a disadvantage given the modern trend towards expanding modulation bandwidths. Further, as systems are driven harder for better power efficiency, they also increase the order of significant distortion products. Linearizing higher degrees of nonlinearity (5ths and 7ths), rapidly expands the bandwidth the DPD must address to several multiples of the fundamentals. Note that these bandwidth and power issues involve not only the DPD but also the digital-to-analog converter (DAC) and the input stages of the upconverter. The power consumed by the digital signal processing subsystem in this situation rapidly diminishes overall system power efficiency to a point of diminishing returns.
Another approach to linearization is “out-of-band” (OOB) linearization. OOB linearization takes advantage of the fact that when an original RF fundamental signal is first multiplied by itself (constituting a second-order operation) as an intermediate step, and that result is again multiplied by the original RF fundamental signal (another second-order operation), then the final result includes third-order products and, importantly, the intermediate step of this cascade-of-multiplications (COM) process constitutes an opportunity to manipulate the vector attributes of the signal at frequencies (either baseband or second harmonic) far removed from, and thus not interfering with, the fundamental RF frequencies. The latter step of the COM process frequency-translates the vector-manipulated signal back to and near the fundamental RF frequencies for desirably destructive interference with the DMA's direct third-order distortion products.
Linearizers based on OOB COM can be based on either a down-then-up (DTU) sequence or up-then-down (UTD) sequence. The UTD sequence is relatively rare because of difficulties of signal processing at the very high second harmonic frequencies and because the load second harmonic impedance is often engineered to be short for efficiency reasons [Cripps]. The more common DTU sequence offers the relative ease of signal processing at the low baseband frequencies and provides an opportunity to address long-term memory effects [Vuolevi].
An example of a DTU COM OOB predistorter is found in U.S. Pat. No. 6,750,709, to Raghavan. The second-degree nonlinearity of a RF input diode power detector generates a downconverted envelope signal at baseband frequencies (the first COM step). Two independently scaled versions of this envelope signal then amplitude modulate the I and Q phases of the original RF signal (the second COM step). This basic concept is extended in U.S. Pat. Appl. Pub. No. 20150127995, by Domokos, and U.S. Pat. No. 6,757,338, to Kim, which teach higher even orders of baseband nonlinear processing to address (after upconversion) fifth and higher degrees of RF nonlinearity.
Another OOB linearization example is the integrated DTU system of Leung and Larson [Leung, Vincent. W; and Larson, Larry, “Analysis of Envelope Signal Injection for Improvement of RF Amplifier Intermodulation Distortion,” IEEE J. Solid-State Circuits, Vol. 40, No. 9, September 2005]. This article teaches an on-chip detector which creates the baseband-frequency envelope of the RF input signal (the first COM step), a scaled version of which dynamically adjusts the bias point of the nearby power amplifier transistor (the second COM step). Lacking vector tunability however, this scalar approach provides only partial linearization and relies on the short time delays relative to a RF period achieved with tight on-chip integration. Disadvantages of DTU OOB COM systems are susceptibility to noise in the RF input envelope detection process and the vagaries of low frequency circuits such as bandwidth limitations and 1/f noise. An RF quadrature phase shifter is also required in the approaches that include vector tuning.
The derivative superposition approach linearizes a FET-based DMA with an added auxiliary parallel FET, sized and biased to embody a compensating transfer function nonlinearity. In an exemplary circuit, the auxiliary FET's gate voltage is biased towards pinchoff, creating relatively little fundamental output but an expansive transfer function characteristic that compensates the compressive transfer function characteristic of the MDA FET. IMD3 products are thereby cancelled. [See, Webster et. al., “Control of Circuit Distortion by the Derivative Superposition Method”, IEEE Microwave and Guided Wave Letters, Vol. 6, No. 3, March 1996.]
Derivative superposition minimizes distortion at the circuit level and so is nicely amenable to IC technology, taking advantage of the tight matching, precise scaling, close physical proximity, and thus good tracking of the main and auxiliary devices. The option also exists to add additional parallel FETs to address yet higher degrees of nonlinearity. Derivative superposition addresses transfer function nonlinearities but does not address the products of the nonlinear output current divide ratio between the nonlinear part of the FET's output admittance and the load. Worse, the nonlinear part of the FET's output admittance contains both real and imaginary parts at RF such that the output distortion product vectors are not necessarily aligned in phase with those from the transfer function nonlinearities. An auxiliary FET bias adjustment, in essence a scalar tuning control, cannot provide compensation.
The vector nature of the distortion signals is addressed in the modified derivative superposition (MDS) approach of Aparin and Larson, wherein a tapped inductor in the FETs' source connections aligns the nonlinear distortion product vectors for proper cancellation. [Aparin, V. and Larson, L. E., “Modified Derivative Superposition Method for Linearizing FET Low Noise Amplifiers”, IEEE Trans. Microwave Theory and Techniques, Vol. 53, No. 2, February 2015, pp 571-581]. Unfortunately the design analysis equations are very complicated, even when considering only a subset of FET and circuit properties. The tapped inductor is also a difficult and physically large design. Further, the design works at only one center frequency and is thus not amenable to multi-band radios. The lack of vector tenability, the intractably difficult analysis, and the insufficiently accurate simulation models for very high levels of distortion suppression mean the design is substantially empirical, requiring several iterations. Even then, process variations limit the statistically available linearity [Ganesan, et. al., “A Highly Linear Low-Noise Amplifier”, IEEE Trans. Microwave Theory and Technique, Vol. 54, No 12, December 2016, pp. 4079-4085].
Another disadvantage of the derivative superposition approach is that it cannot accommodate bipolar junction transistor (BJT) single-ended structures. While the sign of a FET's third-order power series coefficient can be altered with a bias change, a BJT's transfer function derivatives all have the same sign, independent of bias.
The approaches of Garuts and of Kim et al each linearize a BJT main differential transconductance stage with a parallel, lightly biased (and thus overdriven) auxiliary differential transconductance stage and subtractive output summations. The auxiliary stage operates at a very high fractional state of nonlinearity relative to the main transconductance stage such that its distortion product magnitudes equal those of the main stage, but with relatively little (and thus substantially noninterfering) fundamental output content. Both incorporate judicious placement of additional passive elements to fine-tune the high frequency distortion vectors for enhanced cancellation. [Garuts, U.S. Pat. No. 4,835,488] and [Kim, W., et. Al, “A Mixer With Third-Order Nonlinearity Cancellation Technique for CDMA Applications”, IEEE Microwave and Wireless Components Letters, Vol. 17, No. 1, January 2017, pp 76-78] Garuts's low pass design adds resistors to the auxiliary stage bases that effectively gyrate to an equivalent emitter inductance at frequencies above the device fb, maneuvering the auxiliary stage's distortion vectors in the desired manner. Kim et al use degeneration resistors and collector RC phase-shift networks in the auxiliary stage.
U.S. Pat. No. 7,088,980 teaches use of a mixer (“frequency converter”) within a structure similar to Kim et al, and adds a phase-shifter in the auxiliary mixer's local oscillator input instead of the collector RC phase-shift network. [See also, S. Otaka, M. Ashida, M. Ishii, and T. Itakura, “A +10 dBm IIP3 SiGe mixer with IMD3 cancellation technique,” in IEEE ISSC Tech. Dig., February 2004, pp. 398-399.] Otaka also teaches both single-ended and FET-based variants of the transconductance stages. Like derivative superposition, the integrated linearizers of Garuts, Kim et al, and Otaka suffer a difficult design process and high sensitivity to process variations.
In general, maintaining aggressive linearization goals requires compensation for drift over time, voltage, and temperature. Negative feedback reduces drift. Linearizers with local RF negative feedback suffer other problems mentioned previously. Open loop systems such as feedforward and predistortion require adaptivity as a defense against drift. Adaptivity is in essence a larger negative feedback loop around the linearizer/DMA combination. Adaptive systems require a tunable linearizer, and adaptive systems at RF and microwave frequencies require a vector-tunable linearizer. The complexity of an adaptive system is eased considerably with a linearizer embodying orthogonality in its vector tuning controls (meaning controls that do not interact with each other), that is, orthogonal magnitude and phase controls, or orthogonal real part and imaginary part controls.
The foregoing patents reflect the current state of the art of which the present inventor is aware. Reference to, and discussion of, these patents is intended to aid in discharging Applicant's acknowledged duty of candor in disclosing information that may be relevant to the examination of claims to the present invention. However, it is respectfully submitted that none of the above-indicated patents disclose, teach, suggest, show, or otherwise render obvious, either singly or when considered in combination, the invention described and claimed herein.